Integrated, digitally-controlled crystal oscillator

ABSTRACT

A quartz crystal oscillator comprises a balanced circuit with a quartz crystal resonator device connected in series resonance across a balanced, low-impedance node within a sustaining amplifier. A phase modulator such as a quadrature modulator is included in the feedback loop to allow programming of the loop phase shift thereby to alter the frequency point on the crystal resonance curve at which the circuit oscillates. The in-phase loop signal is hardlimited while the quadrature loop signal component is not hardlimited with the effect that the frequency control curve slope is more accurately defined. An active neutralization of the crystal&#39;s parasitic shunt capacitance is disclosed for obtaining a linear frequency control curve.

BACKGROUND OF THE INVENTION

[0001] The present invention relates generally to the field of variablecrystal oscillators and specifically to a variable crystal oscillatorsuitable for inclusion on an integrated circuit.

[0002] Radio frequency mobile communications terminals, such as forexample cellular telephones, require an on-board frequency reference forestablishing operating channel frequencies. The mobile terminal receivesa base station signal and measures the apparent frequency error of thereceived signal. Since the base stations use highly accurate frequencysources, any error in a received signal is attributed to the mobileterminal's own frequency reference. Mobile terminals typically utilizecrystal oscillator circuits whose frequency is electronicallyadjustable. Such circuits are known as Voltage Controlled XtalOscillators (VCXO). The mobile terminal, upon detecting an apparentfrequency error in a received signal, generates a frequency correctionsignal and adjusts the crystal oscillator to reduce the apparentfrequency error. The mobile terminal's reference oscillator accuracy isthereby adjusted to equal that of the base station by automaticfrequency correction (AFC). It is well known in the prior art that themobile terminal may compute the frequency error digitally, the digitalerror value then being applied to a digital to analog (D/A) converter togenerate a correcting voltage to the VCXO.

[0003] Prior art VCXOs are constructed by connecting a quartz crystalresonator in the feedback loop of a sustaining amplifier. Avariable-capacitance diode (Varicap or Varactor diode) is associated tothe crystal circuit to allow frequency adjustment by applying a voltageto the varactor diode to change its capacitance, and thereby to changethe resonant frequency of the circuit formed by the crystal plus thevaractor diode.

[0004] Great strides have been made in reducing the cost of cellularphones by integrating electrical and logical circuits and functions intosilicon or Gallium Arsenide integrated circuits, or chips. However,components such as varactor diodes used in a prior art VCXOs aredifficult to integrate, as they require different semiconductorprocessing steps than the rest of an integrated circuit.

[0005] It is known in the art to vary the frequency of a crystaloscillator by varying a phase shift with the feedback loop, rather thanusing a varactor diode. Furthermore, it is known to produce the varyingphase shift by varying the magnitude of a quadrature signal component.However, prior art VCXOs utilizing this approach have variousdeficiencies. Some do not operate the crystal oscillator in seriesresonant mode. Others implement control circuits with the undesirablecharacteristic that the loop gain is dependent on the quadrature signalcomponent that is varied to alter the oscillator's frequency. This isproblematic, particularly when the VCXO must be controlled over a widefrequency range. Furthermore, if the crystal is connected in arelatively high impedance circuit, the Q-factor at series resonance isreduced, thereby degrading the stability and phase noise of theoscillator and rendering it unsuitable for applications in which thecrystal frequency must be multiplied up to several GHz, as in modernmobile terminals.

[0006] Another known problem with prior art varactor-less VCXOs is thatthe linearity of the frequency control curve is adversely affected bythe crystal's parasitic shunt capacitance. Some prior art solutionscompensate for the crystal parasitic capacitance by employing a variableinductor to linearize the control curve. However, such a component isnot suitable for integration.

[0007] Still other prior art VCXOs maintain a constant gain undervarying quadrature signal components that alter the output frequency,but exhibit a control curve slope that is inversely proportional to theequivalent resistance of the crystal, which is an ill-defined parameter,rendering reliable control difficult.

[0008] When integrating the crystal oscillator function with otherfunctions in a few chips to reduce cost, care must be taken to avoidmutual interference between functions, which can occur as signals go inand out of pins on the chip, relative to a common ground. Thereforebalanced, or differential, circuits are preferred for reducing unwantedcoupling, as is known in the art. Moreover, the crystal in a mobileterminal must be operated at a high Q-factor to obtain low phase noiseafter multiplying the frequency to the 2 GHz range.

SUMMARY OF THE INVENTION

[0009] A quartz crystal oscillator comprises a balanced circuit with aquartz crystal resonator device connected in series resonance across abalanced, low-impedance node within a sustaining amplifier. A phasemodulator such as a quadrature modulator is included in the feedbackloop to allow programming of the loop phase shift thereby to alter thefrequency point on the crystal resonance curve at which the circuitoscillates. The in-phase loop signal is hardlimited while the quadratureloop signal component is not hardlimited with the effect that thefrequency control curve slope is more accurately defined. An activeneutralization of the crystal's parasitic shunt capacitance is disclosedfor obtaining a linear frequency control curve.

BRIEF DESCRIPTION OF DRAWINGS

[0010]FIG. 1 depicts a variety of prior art crystal oscillator circuits;

[0011]FIG. 2 is an idealized equivalent circuit of a parallel resonantcrystal oscillator;

[0012]FIG. 3 is a plot of frequency vs. capacitance for the circuit ofFIG. 3A;

[0013]FIG. 3A is a crystal equivalent circuit;

[0014]FIG. 4 is a plot of the minimum transcondunctance G_(m) vs.capacitance;

[0015]FIGS. 5A, 5B, and 5D are various circuits employing a variablephase shifter to alter the output frequency of a crystal oscillatorcircuit;

[0016]FIG. 5C is a balanced, differential version of the circuit of FIG.5B;

[0017]FIG. 6 is a plot of VCXO frequency vs. transreactance for thecircuit of FIG. 3A;

[0018]FIG. 7 is a plot of transresistance G_(r) vs. frequency;

[0019]FIG. 8 is a plot of transresistance G_(r) vs. frequency when C₀ isneglected;

[0020]FIG. 9 is a plot of transreactance G_(i) vs. frequency when C₀ isneglected;

[0021]FIG. 10A is a block diagram of a VXO circuit;

[0022]FIG. 10B is a circuit diagram of the circuit of FIG. 10A;

[0023]FIG. 10C is a block diagram of an alternate topology of thecircuit of FIG. 10A;

[0024]FIG. 11 is a compound transistor amplifier circuit;

[0025]FIG. 12 is a circuit neutralizing the effect of C₀;

[0026]FIG. 13 is a circuit using a compound transistor amplifier and C₀neutralization; and

[0027]FIG. 14 is a circuit for controlling a balanced modulator with aΣ-Δ input.

DETAILED DESCRIPTION OF THE INVENTION

[0028]FIG. 1 shows various prior art crystal oscillator circuits. TheColpitts oscillator 10 and the Pierce oscillator 12 differ only in whichpoint of the circuit is grounded, and how bias is applied to the activedevice. They both use the crystal 14 in a parallel resonant(high-impedance) mode forming a π-circuit with the crystal 14 as theseries element and capacitors 16 as the shunt elements. The circuitoscillates at the frequency where the π-circuit has a 180-degree phaseshift, where the crystal impedance is inductive between itsseries-resonant frequency and its parallel self-resonant frequency.Since the parallel-resonant crystal oscillator circuits arehigh-impedance, the sustaining amplifier needs to have a high input andoutput impedance, ideally infinite input impedance (or a purecapacitance that can be regarded as part of C1) and a current sourceoutput. These characteristics were fulfilled by vacuum tubes, and aresubstantially fulfilled by the more modern Field Effect Transistor (FET)18.

[0029] The Butler oscillator 20 uses the crystal 14 in series resonant(low impedance) mode, as does the overtone oscillator 22. The circuits20, 22 oscillate at the frequency where the crystal impedance becomeslow and resistive, and the amplifier in this case has zero phase shift,modulo-2π. The overtone oscillator 22 can employ an LC tuned circuit 24to select operation around an overtone or harmonic of the crystal'sfundamental frequency.

[0030] All the circuits of FIG. 1 can be categorized as either using thecrystal 14 in a parallel resonant mode or in a series resonant mode. Inthe parallel resonant mode, the crystal 14 usually acts like theinductor in a resection with shunt capacitors 16 to create a 180-degreephase shift at resonance. The sustaining amplifier is then a phaseinverting amplifier to produce a loop phase shift of zero, modulo-2π.

[0031] In the series resonant mode, the crystal 14 resonates to a lowimpedance and is connected in such a way as to maximize the loop gain atseries resonance. The sustaining amplifier in that case usually has zeroloop phase shift. Because the crystal 14 is used in its low-impedanceresonance mode, the amplifier should have low input and outputimpedances; ideally zero input impedance and a voltage source output.

[0032] A VCXO is usually constructed with the crystal 14 used inparallel resonance, with one or both of the capacitors 16 being avaractor diode, there being no such components to vary in the case ofthe series-resonance oscillators. However, in the present invention,series resonance is a preferred implementation and novel means aredisclosed to vary the frequency of oscillation according to awell-defined control characteristic curve.

[0033]FIG. 2 shows the idealized circuit of a crystal oscillator 30using the crystal 14 in parallel resonance. It is shown in FIG. 2 thatthe necessary condition for oscillation occurs when the crystalreactance X resonates with the series combination of C₁ and C₂, i.e.,when $\begin{matrix}{X = {\frac{1}{{wC}_{1}} + \frac{1}{{wC}_{2}}}} & (1)\end{matrix}$

[0034] The transconductance G_(m) of the amplifier must then be at leastequal to w²C₁C₂R to sustain oscillation.

[0035] The minimum gain requirement is with C₁=C₂=C₀, and a variablefrequency crystal oscillator (VXO) is produced by varying C₀. Since thecrystal reactance X is a function of frequency X(w), equation (1) is atranscendental equation for the frequency w as a function of C₀. It issimpler to choose w and calculate C₀ and thus plot C₀ versus w, as shownin FIG. 3.

[0036] The curve of FIG. 3 was calculated using the crystal equivalentcircuit parameters shown in FIG. 3A. With the parameters of FIG. 3A andthe range of tuning capacitance from 4 pF to 100 pF, FIG. 3 illustratesthat a range of frequency adjustment of +20 KHz to −6 KHz can beachieved. The type of VCXO exemplified by FIG. 3 is not ideal for theseveral reasons.

[0037] If a large tuning range is indeed required, the frequency versuscontrol parameter range is very non-linear. On the other hand, if it isonly required adjust the frequency to compensate for crystal toleranceand temperature effects, which together may total only ±10 ppm (±130Hz), then the VCXO sensitivity of 30 ppm per picofarad around thenominal frequency is too sensitive to the tuning capacitor tolerance.

[0038] In addition, the transconductance gain of the amplifier needed tosustain oscillation varies over a wide range, as shown in FIG. 4.

[0039] To solve the above problems with the prior art parallel resonantVCXOs, the present invention uses the crystal in series resonant modewith a variable phase shifter to adjust the frequency. Various circuitsfor series-resonant crystal oscillators are shown in FIG. 5, which alsosuggests suitable places within the loop that such a phase shifter maybe located. Some aspects of these circuits are known from the prior art.

[0040]FIG. 5A illustrates a circuit using operational amplifiernotation, in which the amplifier has a flat frequency response. Theoperational amplifier 32 has a virtual ground input, i.e., a very lowinput impedance. The operational amplifier 32 effectively causes the ACcrystal current also to flow through its feedback resistor 33,generating an AC output voltage proportional to the crystal current, butphase inverted. The operational amplifier 34 is merely a phase-invertingamplifier to provide the non-inverting overall gain required foroscillation, and has a low output impedance for driving the crystal inseries resonant mode. A variable phase shifter for adjusting operatingfrequency may be located between operational amplifiers 32 and 34.

[0041] In FIG. 5A, assuming the crystal 14 is a series LCR resonator andignoring for the moment its shunt capacitance C₀ of typically 2.8 pF,the crystal impedance would be a minimum at series resonance and equalto the equivalent series resistance (ESR) of exemplary 8.7 Ω. When thefrequency is adjusted to either side of series resonance, the crystalimpedance is of the form R+jX(w) where X is a reactive part that is afunction of frequency w and R is a constant equal to the ESR when C₀ isignored. The loop gain and the AC crystal current would then be reducedin proportion to 1/(R+jX(w)). In order to maintain required loop phaseshift of zero, the phase shifter would need to cause a phase shift ofφ=tan⁻¹(X/R) and the loop gain would need to be increased in magnitudeby the factor {square root}{square root over (R²+X²)}.

[0042] A quadrature modulator may be used to control both gain and phaseshift of a signal. A quadrature modulator comprises a first balancedmodulator to vary the amount of output signal that is in phase with theinput signal to the quadrature modulator (i.e., the real part r) and asecond balanced modulator to vary the amount of output signal that is inquadrature with the input signal (i.e., the imaginary part i). Thequadrature component that is varied by the second balanced modulator maybe created by passing the input signal through a fixed 90-degree phaseshift network.

[0043] Both the phase and the gain variation required may thus beachieved by controlling only the quadrature part of the gain G_(l) ofthe total gain G_(r)+jG_(l) such that$\frac{G_{r} + {jG}_{i}}{R + {jX}}$

[0044] is always real, i.e., $\frac{G_{i}}{X} = \frac{G_{r}}{R}$

[0045] The in-phase gain G_(r), with G_(l) zero, is set to be sufficientfor the circuit to oscillate with the crystal operating at its seriesresonant frequency, where its impedance is real and nominally a minimum.G_(r) thus represents a minimum gain. Varying G_(l) in either directionfrom zero will then cause the frequency w to change so that the crystalreactance ${X(w)} = \frac{G_{i}R}{G_{r}}$

[0046] and will increase the modulus of the gain so that the loop gainis constant. Thus only half of a quadrature modulator is required—thathalf which modulates the quadrature or imaginary part of the loop gainG_(l). The in-phase or real part of the loop gain G_(r) need not bevaried, at least under the assumption that the effect of C₀ can beneglected. The validity of this assumption may be tested by plotting thereal and imaginary gain components required for oscillation versus thefrequency of oscillation, using the actual crystal equivalent circuit ofFIG. 3A.

[0047]FIG. 6 shows the frequency of oscillation versus the quadraturegain, which is the controlling parameter. The gain has the dimensions ofa transimpedance, and is plotted in units of Ohms. The phase of thetransimpedance is equal to${\tan^{- 1}\left( \frac{G_{i}}{G_{r}} \right)};$

[0048] since the quadrature transimpedance G_(l) ranges over much morethan the G_(r)(=ESR) of 8.7 Ω, the phase is near + or −90 degrees overmuch of the plotted range, and the frequency versus phase function is avery non-linear curve over a wide frequency range. On the other hand,the curve of frequency versus the quadrature part of the transimpedanceis more linear, as shown in FIG. 6. This illustrates that it ispreferable to control frequency by varying the quadrature part of thetransimpedance of the sustaining amplifier, i.e. the transreactance,rather than to vary a phase shift, when a large frequency variation isdesired.

[0049]FIG. 7 shows the required in-phase part of the amplifiertransimpedance G_(r) to sustain oscillation, versus frequency offset. Bycomparing with FIG. 4 it is clear that this varies over a much smallerrange for the series resonant VCXO than the transconductance requiredfor the parallel resonant VCXO.

[0050] The required transresistance would be a constant equal to the 8.7Ω ESR were it not for the influence of the crystal shunt capacitance C₀of 2.8 pF, as is shown in FIG. 8, which was computed for C₀=0.0.Likewise, the curve of frequency versus transreactance for C₀=0.0 ismore linear, as shown in FIG. 9. Therefore there is a motivation toattempt to neutralize or compensate for the effect of C₀, as furtherdisclosed below.

[0051] Referring to FIG. 5B, the uncertainty regarding the precisetransresistance required for oscillation is handled by the use of alimiting amplifier between the two integrators. The operation of FIG. 5Bis as follows: The output of limiter 46 is a square wave, which drivesthe integrator operational amplifier 48 using R 49 and C₂ 50 to form atriangular wave. Since the limiter 46 output is a constant amplitudesquare wave, the triangular waveform is also of constant amplitude. Thetriangular wave containing principally a fundamental component is thenfiltered by the crystal impedance that emphasizes the fundamentalcomponent relative to the harmonics, thereby allowing substantially onlya fundamental sine wave current component to flow from the crystal 14.The sine wave current is then integrated by operational amplifier 40 andC₁ 42 to produce the sine wave at the input to phase shifter 44. Thephase shifter 44 produces a phase shifted sine wave to drive the limiter46. The limiter 46 drive voltage depends on the rather impreciselyspecified ESR of the crystal, indicated as the dotted-line resistor 52,but despite variations in the input amplitude, the output of limiter 46remains a constant amplitude signal, thus ensuring that oscillation issustained despite variations in the crystal 14 impedance.

[0052] An improved version of the circuit of FIG. 5B, using balancedcomponents and a differential loop signal, is depicted in FIG. 5C.Balanced amplifier 60 includes two transistors 60A and 60B in apush-pull, or balanced, configuration. The crystal 14 is connectedbetween the emitters of transistors 60A, 60B such that the differentialpair has gain inversely proportional to the crystal impedance, andmaximum gain occurs at crystal series resonance. The fully balancedcircuit minimizes unwanted coupling with other circuit functions thatmay be integrated on to the same chip, thus providing improved noiseimmunity over the corresponding configuration of FIG. 5B. Thedifferential output signal passes through a first balanced integrator62, and then through a phase shifter 64 (with a control input K thatcontrols the degree of phase shift induced). The differential output ofthe phase shifter 64 feeds a balanced hard limiter 66, which generates asquare wave that preserves the phase information, but the which is of asignal amplitude substantially independent of its input signalamplitude. The balanced limiter 66 thus imposes a constant loop gain onthe in-line component of the feedback signal, while preserving the phaseshifter 64 alteration of the quadrature gain. The differential output ofthe balanced limiter 66 feeds a second balanced integrator 68, whichgenerates a triangular waveform, which is filtered by the crystal 14 toemphasize its fundamental frequency component, generating a sine waveoutput of the balanced amplifier 60. This operation is substantiallysimilar to that described above with reference to FIG. 5B, with theaddition of balanced components and a differential feedback signal toreduce coupling and improve noise and crosstalk immunity. Note that thebalanced integrators 62, 68 each contribute 90-degrees of phase shift,and the balanced limiter 66 contributes 180-degrees of phase shift, fora nominal loop phase shift of 360 degrees, or zero phase shift, asrequired for oscillation.

[0053]FIG. 5D depicts the use of differentiators to produce the phaseshift required for oscillation, with the phase shifter 44 altering theoscillator output frequency. The operation of FIG. 5D is essentially thesame as FIG. 5B except that the differentiators produce phase advancesof 90 degrees instead of the phase lags of 90 degrees produced byintegrators. However, differentiators also produce higher gain atovertone frequencies than at the fundamental, whereas integratorsproduce lower gain at overtones than at the fundamental. Thereforeintegrators are preferred when overtone suppression is desired. Theintegrators may be leaky integrators to compensate for incidentaladditional phase lags in practical circuits, and to prevent drift of theDC operating point.

[0054] In both the circuits of FIG. 5B and FIG. 5C, the required phaseshift to produce a given frequency shift is${\tan^{- 1}\left( \frac{R_{i}}{R_{r}} \right)},$

[0055] where R_(r)+jR_(l) is the crystal complex impedance. Thus theslope of frequency versus phase depends on G_(r). If the phase shifter44 or 64 comprises a constant in-phase gain G_(r) and a controlledquadrature gain G_(i), then$G_{i} = {\left( \frac{G_{r}}{R_{r}} \right)R_{i}}$

[0056] shows that the quadrature gain to produce a given R_(i) andtherefore frequency offset also depends on the ESR, R_(r). To avoid thefrequency control sensitivity depending on the ill-specified ESR, thereis therefore a motivation for devising additional circuit improvementsthat will make the control sensitivity largely independent of the ESR.In the prior art, this may have been inadvertently achieved by dilutingthe crystal ESR with a large external resistance. However, this lowersthe Q of operation, which raises undesired phase noise levelssubstantially.

[0057]FIG. 10A depicts a block diagram of one embodiment of a variablecrystal oscillator (VXO) according to the present invention, in whichthe slope of the frequency control curve is substantially independent ofthe crystal ESR. This is accomplished by defining two feedback loops.One includes a limiter that fixes the real gain at a constant andprovides a zero loop phase shift, modulo 360-degrees, to produceoscillation. The other loop is not modulated, and includes a balancedmodulator to vary the quadrature gain and thus alter the oscillatorfrequency. The two feedback loops differ by 90-degrees in phase shift.

[0058] Referring to FIG. 10A, balanced amplifier 70 includes the crystal14 in series resonant mode, connected to low impedance emitters of abalanced (push/pull) transistor amplifier. The differential output ofbalanced amplifier 70 travels in a first feedback loop to balancedintegrator 72, balanced limiter 74, and balanced integrator 76. Theintegrators 72, 76 each contribute a 90-degree phase shift, and thelimiter 74 contributes a 180-degree phase shift, for a loop gain of 360,or zero phase shift, to support oscillation. The gain G_(r) of the firstloop is constant due to the limiter 74 removing any loop amplitudevariation of the differential feedback signal.

[0059] The circuit of FIG. 10A includes a second feedback loop, whereinthe differential outputs of balanced amplifier 70 are connected to abalanced modulator 78. Modulator 78 is operative to alter the gain G_(r)of the second loop in response to differential control input K. Thesecond loop differential signal then joins the first loop signal atsumming junction 79, and passes through the second balanced integrator76. The second loop phase shift is 90 degrees (induced by integrator76), and is thus 90 degrees out of phase with the first loop signal.Variations in the non-limited second loop signal are thus in quadratureto the first loop signal; in this manner the modulator 78 may becontrolled to alter the oscillator output frequency.

[0060] A transistor-level integrated circuit embodiment of the VXO ofFIG. 10A is illustrated in FIG. 10B. A differential pair of transistorsTR1A, TR1B driven by emitter followers TR2A, TR2B and with collectorloads TR3A, TR3B forms a push-pull, or balanced, amplifier. The crystal14 is connected between the emitters of TR1A and TR1B such that thedifferential pair has gain inversely proportional to the crystalimpedance, and maximum gain occurs at crystal series resonance. Thefully balanced circuit minimizes unwanted coupling with other circuitfunctions that may be integrated on to the same chip.

[0061] The output of the differential input stage is passed throughemitter followers TR4A, TR4B to a Gilbert-cell balanced modulator TR10A,TR10B, TR10C, TR10D. The collector load for the balanced modulator isformed from TR9A, TR9B and has a high impedance 2R to the differentialmode, and a low impedance R/2·Beta to the common mode. The differentialmode load impedance is sufficiently high that C₁ dominates and forms anintegrator at the modulator output.

[0062] The same signal that is fed to the balanced modulatoradditionally feeds a second integrator TR6A, TR6B with C₂ as theintegrating capacitor. The output of the second integrator is limited bydifferential limiter tr7a, tr7b, tr8a, tr8b and the resultingsquare-wave current of constant magnitude is fed to the first integratorC₁. The output of the first integrator C₁ is fed back to drive thedifferential input amplifier TR1A, TR1B. Thus there is a first feedbackloop around the crystal comprising an integrated-limited-integratedsignal, the two integrators having together a 180-degree phase shiftthat is turned to a zero loop phase shift by selecting the differentialinput connections appropriately. This loop alone will cause oscillationat the frequency where the crystal impedance is real.

[0063] The balanced modulator adds a non-limited contribution to thefeedback signal in a second feedback loop which is only integrated once,by C₁, and is thus in quadrature with the doubly integrated and limitedfirst feedback loop. Moreover, the quadrature feedback loop is notlimited. The quadrature feedback contribution may be varied in magnitudefrom negative through zero to positive by controlling the currentsources in its tail circuit to provide more current to one side or theother of the modulator, thereby causing the oscillation frequency toshift around the crystal series resonance point.

[0064] The circuit of FIGS. 10A, 10B has the advantage that limiting thein-phase feedback path and not limiting the quadrature feedback pathmakes the slope of the frequency control curve independent of thecrystal 14 ESR. The completely balanced circuit is also advantageous forintegration on a chip used in a sensitive radio transmitter-receiversuch as a mobile terminal, while avoiding interference due to unwantedradiation or coupling of the crystal frequency to other circuits.

[0065] Both results are additionally achieved in the circuit depicted inblock diagram form in FIG. 10C. 10C depicts an alternate topology of thecircuit of FIG. 10A, and like components are numbered correspondingly.FIG. 10C depicts the same two feedback loops: a limited loop with zeronet phase shift formed by integrator 72, limiter 74, and integrator 76;and a non-limited loop with 90-degree phase shift formed by integrator72, and including a modulator 78. The two loops are joined at summingjunction 79 and feed back to balanced amplifier 70, which includes thecrystal 14 in series resonance mode between the low impedance emittersof a push-pull transistor amplifier stage. Operation of the oscillatorcircuit of FIG. 10C is directly analogous to that described aboverelating to FIGS. 10A and 10B.

[0066] Referring again to FIG. 10B, the gain of the differential inputstage may be limited by the TR1A, TR1B emitter-to-emitter impedance of50/I_(e), which can be higher than the crystal 14 ESR for small I_(e).It is desirable to conserve current I_(e) in battery-operatedapplications. Therefore either a higher impedance crystal havinginductance in the 50 mH region and ESR in the 100 Ω region might beused, or else TR1A, TR1B can be replaced by a compound transistoramplifier. FIG. 11 depicts such a compound transistor amplifier, formedby TR1A, TR1C, and TR1B, TR1D. The emitter-to-emitter impedance of thecompound transistor stage is reduced by the Beta of TR1C, TR1D to$\frac{50}{{Beta} \cdot I_{e}}.$

[0067] For Beta=50 and I_(e)=1 mA, this gives 1.0 Ω which is now smallcompared to the 8.7 Ω ESR of the exemplary crystal 14. The objective ofoperating the crystal at maximum Q is realized when substantially theonly resistance in series with the crystal 14 is its own ESR. However,the control curve slope would, without the use of a limiter, then bedependent on the ill-defined ESR. The use of the limiter for thein-phase component of the loop feedback signal makes the control curveslope independent of the ESR.

[0068]FIG. 12 depicts, according to one embodiment of the presentinvention, how the influence of the crystal shunt capacitance C₀ may beneutralized. A first differential input transistor pair TR1A, TR1B hasthe crystal 14 connected between its emitters. A second pair TR1E, TR1Fhas only a neutralizing capacitor C_(n) connected between its emitters.The collectors of TR1E, TR1F are then cross-connected to the collectorsof TR1A, TR1B so that the current due to the neutralizing capacitorC_(n) adds in antiphase to the current due to the crystal C₀ and therebycancels. In FIG. 12, TR1A, TR1B, TR1E, TR1F can all be compoundtransistors as depicted in FIG. 11.

[0069]FIG. 13 depicts a complete differential input stage using compoundtransistors with neutralization of C₀. Transistors A, E form onecompound input transistor of a first differential pair and B, F form theother side of the pair. The first differential pair has the crystal 14connected between the emitters. A second differential pair is formed bythe compound transistors C, G and D, H and has a neutralizing capacitorC_(n) connected between its emitters. The compound transistor collectorsfor the first differential pair are the emitters of transistors E and Fand the collectors of the second differential pair are the emitters oftransistors G and H. The latter are connected to the former with areversal so that current due to C_(n) in the second differential pairadds in antiphase to the current in the first differential pair due tothe crystal shunt capacitance Co. The combined collectors have collectorloads RL. When the crystal ESR is as low as 8.7 Ω, an RL of 100 Ω givesa gain of 2×100/8.7=23. The total current through RL may be as low as 1mA so that the DC voltage drop is only 100 mV. Double emitter followersP, Y and Q, Z drop the DC level of the output signal a further 2 Vbe toabout the same level as the circuit of FIG. 10, and therefore canconnect directly to the modulator and second integrator as shownpreviously.

[0070] The circuits of FIGS. 10 and 13 are designed to operate from avoltage as low as 2.7V, which allows operation from a singlerechargeable Lithium cell.

[0071] In the circuit of FIGS. 10A, 10B, 10C, the output signal point isnot expressly depicted. An output may be taken from a variety of points,and its selection is within the discretion of the system designer. Ingeneral, the following points are relevant to the selection of an outputpoint:

[0072] (i) The output signal should be taken from a point in the circuitwhere the signal has been filtered by the frequency selective resonanceof the crystal, and before any stage of the circuit that may degrade thesignal to noise ratio.

[0073] (ii) The output signal should be taken through a buffer stagesuch that uncertainties in the load driven by the output signal changeneither the loop gain nor the loop phase shift, in order to avoid “loadpulling.”

[0074] Those of skill in the art will readily realize that theoscillator circuit of the present invention may be formed using fieldeffect transistors, e.g. CMOS FETs, and that compound transistors mayalso be formed using FETs to achieve low crystal drive impedance.Implementation in the so-called Bi-CMOS, process in which both bipolarjunction transistors and CMOS FETs are available, is additionally withinthe scope of the present invention.

[0075] Often, the tuning range of a VCXO need only be sufficient toadjust the crystal frequency to compensate for an initial cuttingtolerance of ±10 ppm plus a temperature variation of ±10 ppm in order toform a Temperature Compensated Crystal Oscillator or TCXO. A controlsignal may be generated in response to a temperature sensor to provideopen-loop frequency control in the absence of other frequency errorinformation. When a frequency accurate signal is received, a frequencyerror relative to the received signal may be measured and the controlsignal altered to correct the measured error. In prior art mobileterminals, the control signal is generated as a digital signal by asignal processor and converted to an analog control signal by a D/Aconverter for application to, e.g., a varactor diode. In the circuit ofFIG. 10B, the control signal is applied to the current sources in themodulator tail circuit to vary the quadrature feedback ortransreactance. The desired control range of ±20 ppm is only ±260 Hz fora 13 MHz oscillator, which is much smaller than the range of controlplotted in the FIGS. 3, 4, 6, 7, 8 and 9. The control range may bediluted by reducing the amount of transreactance created by the balancedmodulator, by scaling its tail currents and the gain distribution toprovide the range of control desired. The current sources may becontrolled using analog control voltages, preferably from a balancedSigma-Delta D/A converter. The modulator can operate in “class B,” asdisclosed in U.S. Pat. No. ______ to Applicant Dent and co-inventorHadjichristos, which is hereby incorporated by reference in itsentirety. Using the class-B modulator, both current sources arenominally at zero current when no transreactance is desired. When apositive transreactance is desired, the current in one current source isincreased while the other remains nominally zero. When a negativetransreactance is desired, the other current source is increased fromzero while the former is held at zero current. The current sources canbe formed by current mirrors so that their currents reflect controlcurrents from the D/A converter, as described in the above-incorporatedpatent and depicted in FIG. 14. In FIG. 14, the input sigma-deltabitstream turns the P-type current mirrors alternately on and off. TheP-type current is filtered by the filter R-C-R and the filtered currentis mirrored by the N-type current mirrors to provide differentialcontrolled currents I+ and I−.

[0076] Alternatively, the controlled current sources can comprise D/Aconversion, accepting a set of digital control bits that turn on or offa series of paralleled current sources with relative scalings of1:½:¼:⅛. . . In this case, steps to maintain monotonicity of the controlcurve may be needed if more than 256 steps of current are desired, forexample by the use of a coarse and a fine current D/A in parallel, eachdriven by a separate control byte.

[0077] The design of VCXO/TCXO circuits using the techniques disclosedabove enables accurate and stable variable-frequency crystal oscillatorsto be constructed on an integrated circuit with the minimum number ofexternal components, thereby reducing the cost and size of consumerproducts such as mobile terminals.

[0078] Although the present invention has been described herein withrespect to particular features, aspects and embodiments thereof, it willbe apparent that numerous variations, modifications, and otherembodiments are possible within the broad scope of the presentinvention, and accordingly, all variations, modifications andembodiments are to be regarded as being within the scope of theinvention. The present embodiments are therefore to be construed in allaspects as illustrative and not restrictive and all changes comingwithin the meaning and equivalency range of the appended claims areintended to be embraced therein.

What is claimed is:
 1. A variable frequency oscillator circuit,comprising: a resonant component; a sustaining amplifier connected tosaid resonant component and generating an output signal, a firstfeedback loop connected to said sustaining amplifier for an in-phasecomponent of said signal, said first loop including a limiter having anoutput amplitude signal that is substantially independent of its inputamplitude signal; and a second feedback loop connected to saidsustaining amplifier for a quadrature signal component of said signalbypassing said limiter; wherein the output frequency of said oscillatorcircuit is varied by altering said quadrature signal component.
 2. Theoscillator circuit of claim 1 wherein the total loop phase shift iszero, modulo 2π.
 3. The oscillator circuit of claim 2 wherein saidin-phase loop includes two components introducing a 90-degree phaseshift and one component introducing a 180-degree phase shift, and saidquadrature loop includes one component introducing a 90-degree phaseshift.
 4. The oscillator circuit of claim 3 wherein said componentsintroducing a 90-degree phase shift comprise integrators.
 5. Theoscillator circuit of claim 3 wherein said components introducing a90-degree phase shift comprise differentiators.
 6. The oscillatorcircuit of claim 3 wherein said component introducing a 180-degree phaseshift comprises said limiter.
 7. The oscillator circuit of claim 1,further comprising a modulator operative to alter said quadrature signalcomponent in response to a control input, to control the outputfrequency of said oscillator circuit.
 8. The oscillator circuit of claim7 wherein said the gain of said circuit remains constant as saidquadrature signal is altered.
 9. The oscillator circuit of claim 7wherein the relationship between a change in said control input and acorresponding change in said output frequency is substantiallyindependent of the equivalent series resistance of said resonantcomponent.
 10. The oscillator circuit of claim 1 wherein said sustainingamplifier is balanced, and said signal comprises a differential pair.11. The oscillator circuit of claim 1 wherein said resonant component isa piezo-electric device.
 12. The oscillator circuit of claim 11 whereinsaid piezo-electric device is a quartz crystal.
 13. The oscillatorcircuit of claim 1 wherein the nominal resonant frequency of saidresonant component is the series-resonant frequency at which saidresonant component exhibits a minimum impedance.
 14. The oscillatorcircuit of claim 1 wherein said output frequency is varied around theseries resonant frequency of said resonant component.
 15. The oscillatorcircuit of claim 1 further comprising a circuit for neutralizing theinfluence of the shunt self-capacitance of said resonant component. 16.The oscillator circuit of claim 1 wherein the phase shift of said secondloop differs from the phase shift of said first loop by ±90 degrees. 17.The oscillator circuit of claim 1 wherein said sustaining amplifier isbalanced, and said first and second loops comprise differential pairpaths through balanced components.
 18. The oscillator circuit of claim 1wherein one of said first or second loop includes an odd number ofintegrators or differentiators and the other loop includes an evennumber of integrators or differentiators.
 19. The oscillator circuit ofclaim 1 wherein the relative gain of said first and second feedbackloops are controlled to produce a controllable phase shift whilemaintaining a constant loop gain.
 20. The oscillator circuit of claim 19further including a modulator in said first or said second loop, but notboth, to control said relative gain.
 21. The oscillator circuit of claim20 wherein said relative gain varies from a negative value through zeroto a positive value.
 22. The oscillator circuit of claim 20 wherein thephase of the gain of the loop including said modulator contributes aquadrature component to the loop gain having a phase shift equal to ±90degrees.
 23. A variable frequency oscillator circuit, comprising: abalanced, sustaining amplifier having differential inputs and outputs; acrystal oscillator connected across said amplifier; and a feedback loopcomprising a first balanced phase shift component having a differentialinput connected to said amplifier differential output, and adifferential output; a balanced phase shifter having a differentialinput connected to said first phase shift component differential output,a control input operative to alter the amount of phase shift, and adifferential output; a balanced limiter having a differential inputconnected to said phase shifter differential output, and a differentialoutput whose signal amplitude is substantially independent of the signalamplitude of its differential input; and a second balanced phase shiftcomponent having a differential input connected to said limiterdifferential output, and a differential output connected to saidamplifier differential input; and a control signal connected to saidphase shifter control input, whereby the output frequency of saidvariable frequency oscillator circuit varies in response to said controlsignal.
 24. The oscillator circuit of claim 23 wherein said crystaloscillator operates in series resonant mode.
 25. The oscillator circuitof claim 23 wherein said output frequency is varied around the seriesresonant frequency of said crystal oscillator.
 26. The oscillatorcircuit of claim 23 wherein the loop phase shift of said oscillatorcircuit is zero, modulo 360 degrees.
 27. The oscillator circuit of claim23 wherein said first and second balanced phase shift components eachimpart a 90-degree phase shift, and said balanced integrator imparts a180-degree phase shift.
 28. The oscillator circuit of claim 27 whereinsaid first and second balanced phase shift components comprise balancedleaky integrators.
 29. The oscillator circuit of claim 27 wherein saidfirst and second balanced phase shift components comprise balanceddifferentiators.
 30. The oscillator circuit of claim 23 wherein saidbalanced phase shifter comprises a balanced quadrature modulator. 31.The oscillator circuit of claim 23 wherein said balanced limiter isoperative to maintain the amplitude of the feedback signal constant,while passing the varying phase generated by said balanced phaseshifter.
 32. The oscillator circuit of claim 23 wherein the loop gain isconstant.
 33. A variable frequency oscillator circuit having shuntcapacitance neutralization, comprising: a balanced amplifier havingdifferential inputs and outputs, comprising a first pair of transistors,said differential inputs connected to the gates of said transistors andsaid differential outputs comprising the collectors of said transistors;a crystal oscillator having a shunt capacitance connected between theemitters of said balanced amplifier transistors; a second pair oftransistors, the collectors of said second pair cross-connected to thecollectors of said first pair of transistors; and a neutralizingcapacitor connected between the emitters of said second pair oftransistors, whereby current flow through said first pair of transistorsdue to the shunt capacitance of said crystal oscillator is cancelled bycurrent flow through said second pair of transistors due to saidneutralizing capacitor.